This invention is generally in the field of radar and similar ranging techniques for identifying remote targets, and relates to a signal structure to be transmitted towards a remote target which is to be identified.
Radar and sonar systems identify targets and the range of targets by transmitting energy towards the target, and measuring the time between the transmission and reception of an echo from the target. A system of such kind typically comprises such main constructional parts as a signal generator/transmitter, an echo receiver, a filtering means, and a signal processing means. The filtering means typically include a Doppler filter aimed at identifying moving targets and distinguishing among targets moving with different radial velocities.
It is a common goal of such systems to improve the system resolution. Resolution is determined by the relative response of the radar to targets separated from the target to which the radar is matched. In other words, a target is set to be resolved if its signal is separated by the radar from those of other targets in it least one of the coordinates used to describe it.
The high speed and long range of modern airborne vehicles place increasing range demands on radar systems used for tracking. The long-range requirement typically requires relatively high transmitted energy (to detect small targets), which implies a relatively high peak transmitted power or a longer duration transmitter pulse. The latter reduced range resolution, i.e., the ability to distinguish among targets that are at similar ranges.
Pulse compression techniques are known to improve the range resolution in spite of the longer pulse duration. A technique involving frequency dispersion by transmitting a variable frequency xe2x80x9cchirpxe2x80x9d pulse allows the use of pulse compression filters at the receiver to reduce the effective pulse duration to thereby restore range resolution.
The main problem associated with pulse compression is the appearance of range sidelobes in addition to the main range lobe. The time position, or range, of the main lobe is the position that is tested for the presence of a target and for estimating the parameters of that target (i.e., reflected energy or power closing speed, fluctuations in echo power and closing speed, etc.). The presence of range sidelobes on the compressed pulse results in interfering echoes which originate at ranges other than the range of the main lobe. This interference can cause erroneous estimates of the echo characteristics in the range increment covered by the main lobe.
One of the known techniques for suppressing range sidelobes is consists in applying phase coding to the transmitted pulse, so that the coding appears in the received echo pulse, and in applying code-matched filtering to the compressed received pulses.
According to another known technique of the kind specified, complementary phase sequences are imposed on the transmitted signal. This technique is disclosed for example in U.S. Pat. No. 5,151,702. Here, the transmitted pulses are organized into mutually complementary sets. More specifically, pairs of complementary phase sequences are transmitted sequentially, sequentially Doppler filtered, and the filtered pulse sets are, in turn, compressed by filtering matched to the coding. U.S. Pat. No. 5,376,939 discloses a radar system in which transmission takes place simultaneously at two different frequencies, spaced far apart, and in which each of the transmissions is coded with one of two mutually complementary codes.
Generally, xe2x80x9ccomplementary codesxe2x80x9d are basically characterized by the property that the autocorrelation vector sum is zero everywhere except for the zero shift. Two pulses are xe2x80x9cmutually complementaryxe2x80x9d in that, after pulse compression by matched filtering, the sidelobes are equal but of opposite sign, while the main lobes add producing an enhanced main lobe with no sidelobes.
U.S. Pat. No. 5,963,163 discloses a technique for frequency-modulated continuous wave (FMCW) radar detection wit removal of ambiguity between the distance and the speed. According to this technique, the radar sends out at least alternately two parallel and discontinuous frequency modulation ramps that are slightly offset by a frequency variation. The frequency switches from one ramp to the other at the end of a given duration. The distance from a detected target is estimated as a function of the difference in phase between a received signal corresponding to the first ramp and a received signal corresponding to the second ramp. The speed of the target is obtained from the estimated distance and the ambiguity straight line associated with the target.
It is known that range (delay) resolution is inversely related to the radar signal bandwidth. The quest for higher bandwidth usually follows shorter bit duration in digital phase modulated signals, or wider frequency deviation in analog frequency modulated signals. In radio communications, where it is advantageous to increase bit-rate without shortening the bit duration, one solution is the use of a modulation technique known as Orthogonal Frequency Division Multiplexing (OFDM). The main principles and advantageous features of OFDM technique suggested for Digital Audio Broadcasting and other applications are disclosed, for example, in the article xe2x80x9cDigital Sound Broadcasting to Mobile Receiversxe2x80x9d, Le Floch, Halbert-Lassalle, B. R,. and Castelain D., IBEEE Trans. Consum. Elec., 1989, 35, (3) pp. 493-503, and in U.S. Pat. Nos. 5,862,182 and 6,021,165.
OFDM broadcasts have multiple subcarrier signals, on which data are transmitted in parallel. The basic idea of OFDM is to replace transmitting serially M short modulation symbols each of duration tc, by transmitting M long symbols, each of duration tb such that tb=Mtc, wherein these M long modulation symbols are transmitted in parallel on M different subcarriers. In OFDM, the subcarriers are separated by 1/tb, which ensures that the subcarrier frequencies are orthogonal and phase continuity is maintained from one symbol to the next.
As for the radar systems, simultaneous use of several subcarriers there was recently disclosed in the following article: Jankiraman, M., B. J. Wessels, and P. Van Genderen, xe2x80x9cSystem Design and Verification of the PANDORA Multifrequency Radarxe2x80x9d, Proc. of Int""l. Conf On Radar Syst., Brest, France, 17-21 May 1999, Session 1.9. Here, FMCW radar achieves bandwidth of 384 MHZ by using eight Linear-FM (LFM) channels, each sweeping 48 MHz. Together with guard bands, the bandwidth totals 776 MHz. A multifrequency signal is characterized by varying amplitude. Amplifying such a signal requires linear power amplifiers (LPA), which are relatively inefficient. The technique disclosed in this article is directed towards power combining and amplification.
A modern replacement of the analog LFM signal is a digital phase-coded signal, for example the polyphase codes P1, P2, P3 and P4 disclosed in the following publication: Kretschmer, F. F. and Lewis, B. L., xe2x80x9cDoppler Properties of Polyphase Coded Pulse Compression Waveformsxe2x80x9d, IEEE Trans. Aerosp. Electron. Syst., 1983, 19, (4), pp. 521-531. These signals are such that their phase sequences are samples from the phase history of a LFM signal. These codes can be obtained from considering the sampled phases of the step-chirp and chirp baseband waveforms. These codes can be digitally compressed by using fast Fourier transform (FFT) directly or by a fast convolution technique.
The present invention provides a novel multifrequency signal structure for use in the radar or the like target detection system. The main idea of the present invention is based on the inventor""s investigation showing that lower autocorrelation sidelobes are reached when M sequences, modulated on the M subcarriers, are different from each other and constitute a complementary set. The inventor calls such a signal structure as Multifrequency Complementary Phase Coded (MCPC) signal of size Mxc3x97M.
The signal structure according to the invention utilizes M subcarriers simultaneously. The subcarriers are phase modulated by M different sequences that constitute a complementary set. Such a set can be constructed, for example, from the M cyclic shifts of a perfect phase-coded sequence of length M (e.g., P4 signal). The subcarriers are separated by the inverse of the duration of a phase element tb, yielding the above-indicated OFDM feature, well known in communications. The signal exhibits a thumbtack ambiguity function with delay resolution of tb/M.
The ambiguity function |X(xcfx84,v)| is known to be defined as follows:       "LeftBracketingBar"          X      ⁡              (                  τ          ,          v                )              "RightBracketingBar"    =      "LeftBracketingBar"                  ∫                  -          ∞                ∞            ⁢                        u          ⁡                      (            t            )                          ⁢                              u            *                    ⁡                      (                          t              -              τ                        )                          ⁢                  exp          ⁡                      (                          j2π              ⁢                              xe2x80x83                            ⁢              vt                        )                          ⁢                  ⅆ          t                      "RightBracketingBar"  
wherein xcfx84 is the time delay between the reference (stored in the receiver) and received signals; v is Doppler shift; t is the time coordinate; u(t) is complex envelope of the signal; u*(t) is its complex conjugate; and j=(xe2x88x921)xc2xd.
The complex envelope u(t) of a real signal s(t)=g(t)cos[2xcfx80fct+xcfx86(t)], wherein g(t) is real envelope, xcfx86(t) is phase, and fc is carrier frequency, is defined as follows:
u(t)=g(t)cos[xcfx86(t)]+j g(t)sin[xcfx86(t)]
The autocorrelation of u(t) is determined as follows:       r    ⁡          (      τ      )        =            ∫              -        ∞            ∞        ⁢                  u        ⁡                  (          t          )                    ⁢                        u          *                ⁡                  (                      t            -            τ                    )                    ⁢              ⅆ        t            
wherein |r(xcfx84)|=|X(xcfx84,0)|.
The power spectrum is relatively flat, with a width of M/tb. The signal structure can be constructed by power combining M fixed-amplitude signals. The resulting signal, however, is of variable amplitude. The peak-to-mean envelope power ratio can be maintained below 2. A train of complementary pulses and a weight function along the frequency axis can be used for further sidelobe reduction.
Power spectral density Pu(f) of a finite-duration signal with complex envelope u(t) is determined as follows:
xe2x80x83Pu(f)=|U(f)|2
wherein U(f) is the Fourier transform of u(t), that is:       U    ⁡          (      f      )        =            ∫              -        ∞            ∞        ⁢                  u        ⁡                  (          t          )                    ⁢              exp        ⁡                  (                                    -              j2π                        ⁢                          xe2x80x83                        ⁢            f            ⁢                          xe2x80x83                        ⁢            t                    )                    ⁢              ⅆ        t            
There is thus provided according to the present invention a multifrequency phase-coded signal structure to be used in a system for detecting a remote target, the signal structure comprising at least one pulse signal in the form of a mutually complementary set of M sequences, each sequence being composed of M phase-modulated bits, wherein each two adjacent sequences are modulated on subcarriers separated by a frequency fs such that fs=1/tb, tb being a bit duration, and wherein all the subcarriers are transmitted simultaneously.
The term xe2x80x9csignal structurexe2x80x9d utilized herein signifies that the entire number of M sequences within the structure, modulated on the M subcarriers, are transmitted simultaneously as a common pulse signal.
An extension of the xe2x80x9csignal structurexe2x80x9d is a coherent train of M pulses separated in time, wherein each pulse in the train is designed as described above, thus achieving complementary in time as well as in frequency. To this end, for example, each pulse in the train may exhibit a different order of the same complementary set of sequences, such that a set of complementary phase sequences is obtained in each subcarrier frequency. In other words, such a signal structure presents a matrix, whose columns are the pulses separated in time and raws are the subcarrier separated in frequency. The phase elements in each raw and in each column constitute a complementary set.
Frequency weighting can be applied by assigning different amplitude to each subcarrier in the case of the single pulse, or by maintaining, over all the pulses, same amplitude in the subcarriers with the same frequency, in the case of the so-called xe2x80x9cextended structurexe2x80x9d. Different amplitude values may be assigned in accordance with the symmetry condition. This means, that if 5 subcarriers are used, then the amplitude of 1st subcarrier is equal to that of the 5th subcarrier, and the amplitude of the 2nd subcarrier is different from that of the 1st and 5th subcarriers, but equal to that of the 4th subcarrier.
The complementary set may be constructed based on a phase sequence in the form of a polyphase code signal, such as P4 or P3. The phase modulation may also be 2-valued, for example utilizing Golomb code signal, or Ipatov code signal.
The above signal structure can be used in a radar system of any known kind, provided its transmitter unit is capable of simultaneously generating M subcarrier frequencies which define together the multifrequency signal designed as described above, and its receiver is matched to his signal structure.
Generally speaking, when ordering the complementary set over the M frequencies, at least one of the following conditions should be satisfied:
(1) Low autocorrelation sidelobes RMS (root-mean-square or effective value), SLRMS. The RMS of the autocorrelation r(t) sidelobes of an MCPC signal is as follows:       SL    RMS    =                    1                  M          ⁢                      xe2x80x83                    ⁢                      t            b                              ⁢                        ∫                      τ            =                                          t                b                            M                                            M            ⁢                          xe2x80x83                        ⁢                          t              b                                      ⁢                                            [                              r                ⁡                                  (                  τ                  )                                            ]                        2                    ⁢                      ⅆ            τ                              
(2) Low peak autocorrelation sidelobes, SLpeak, which is determined as follows:
SLpeak=max[r(xcfx84)], tb less than |xcfx84| less than Mtb
(3) Low peak sidelobes of the two-dimensional ambiguity function; and
(4) Low peak-to-mean envelope power ratio (PMEPR)